Primary-side controlled switching regulator

ABSTRACT

A switching regulator includes a switching device to switch a transformer from a primary-side to secondary side. A control circuit generates a switching signal for regulating output of the switching regulator. The control circuit includes a first circuit to generate a first signal and a timing signal by measuring a reflected signal of the transformer. A second circuit produces a second signal by integrating a current signal with the timing signal. The current signal represents a primary-side switching current of the transformer. A first feedback circuit produces a first feedback signal in response to the first signal and the reference signal, in which the reference signal is varied in response to the change of the second signal. Furthermore, a second feedback circuit generates a second feedback signal in response to the second signal. A switching control circuit generates the switching signal in response to the feedback signals.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation application of U.S. application Ser. No. 11/270,867, filed on Nov. 8, 2005, which now is U.S. Patent No. 7,352,595. The entirety of the above-mentioned patent application is hereby incorporated by reference therein and made a part of this specification.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a switching regulator, and more specifically it relates to an isolated switching regulator.

2. Description of Related Art

Various switching regulators have been widely used for providing regulated voltage and current. For safety reasons, off-line switching regulators must provide isolation between their primary side and secondary side. Therefore, a control circuit is equipped at the primary side of a switching regulator, an optical-coupler, and the secondary-side circuit are needed to regulate an output voltage and an output current of the switching regulator. To reduce the size and cost of the switching regulator, the need of optical-coupler and secondary-side circuit to be eliminated is the trend.

In recent developments, many primary side control schemes have been disclosed such as U.S. Pat. No. 6,721,192 entitled “PWM controller regulating output voltage and output current in primary side” Yang et al., U.S. Pat. No. 6,836,415 entitled “Primary-side regulated pulse width modulation controller with improved load regulation” Yang, et al; and U.S. Pat. No. 6,862,194 entitled “Flyback power converter having a constant voltage and a constant current output under primary-side PWM control” Yang, et al. However, the drawback of the aforementioned conventional primary side control schemes is the inaccurate control of the output voltage and output current.

The object of the present invention is to provide a switching regulator that is able to accurately control the output voltage and the output current at the primary side of the switching regulator. Therefore, the size and cost of the switching regulator is reduced.

SUMMARY OF THE INVENTION

A primary-side controlled switching regulator (switching regulator) includes a switching device to switch a transformer for transferring the energy from a primary-side to a secondary side of the switching regulator. A control circuit is coupled to the transformer for generating a switching signal to switch the switching device and to regulate the output of the switching regulator. A first circuit coupled to the transformer is included in the control circuit for generating a first signal and a timing signal by measuring a reflected signal of the transformer. A discharge time of the transformer is represented by the timing signal. A second signal is produced by a second circuit and a third circuit by integrating a current signal with the timing signal, in which the current signal represents a primary-side switching current of the transformer. In addition, a time constant of the third circuit is correlated with a switching period of the switching signal. A first error amplifier having a first reference signal is utilized to generate a first feedback signal in response to the first signal. To improve the load regulation, the first reference signal is increased in response to the increase of the second signal. A second feedback signal is generated by a second error amplifier having a second reference signal in response to the second signal. Accordingly, the switching signal is generated by a switching control circuit in response to the first feedback signal and the second feedback signal. The switching signal has a minimum on-time when the switching signal is enabled, which further ensures a minimum value of the discharge time for the multi-sampling of the reflected signal.

It is to be understood that both the foregoing general descriptions and the following detailed descriptions are exemplary, and are intended to provide further explanation of the invention as claimed. Still further objects and advantages will become apparent from a consideration of the ensuing description and drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings are included to provide further understanding of the invention, and are incorporated into and constitute a part of this specification. The drawings illustrate embodiments of the invention and, together with the description, serve to explain the principles of the invention.

FIG. 1 illustrates a conventional circuit diagram of a primary-side controlled switching regulator.

FIG. 2 illustrates a plurality of key waveforms of a conventional switching regulator.

FIG. 3 illustrates a control circuit according to an embodiment of the present invention.

FIG. 4 illustrates a first circuit according to an embodiment of the present invention.

FIG. 5 illustrates an oscillator according to an embodiment of the present invention.

FIG. 6 illustrates a second circuit according to an embodiment of the present invention.

FIG. 7 illustrates a third circuit according to an embodiment of the present invention.

FIG. 8 illustrates a PWM circuit according to an embodiment of the present invention.

FIG. 9 illustrates a circuit schematic of an adder according to an embodiment of the present invention.

FIG. 10 illustrates an adjust circuit for programming a reference signal according to an embodiment of the present invention.

DESCRIPTION OF THE EMBODIMENTS

FIG. 1 illustrates a conventional primary-side controlled switching regulator. The switching regulator includes a transformer 10 having an auxiliary winding N_(A), a primary winding N_(P), and a secondary winding N_(S). To regulate an output voltage V_(O) and an output current I_(O) of the switching regulator, A switching signal V_(PWM) is generated by a control circuit 70 to a transistor 20 for switching a transformer 10.

FIG. 2 illustrates a plurality of signal waveforms of the conventional switching regulator illustrated in FIG. 1. As the switching signal V_(PWM) is a logic-high, a primary-side switching current I_(P) shall be generated accordingly. A peak value I_(P1) of the primary-side switching current I_(P) is given by:

$\begin{matrix} {I_{P\; 1} = {\frac{V_{IN}}{L_{P}} \times T_{ON}}} & (1) \end{matrix}$ where V_(IN) is an input voltage applied to the transformer 10, L_(P) is the inductance of the primary winding N_(P) of the transformer 10, and T_(ON) is an on-time of the switching signal V_(PWM).

Once the switching signal V_(PWM) is dropped to a logic-low, the energy stored in the transformer 10 shall be delivered to the secondary side of the transformer 10 and to an output of the switching regulator via a rectifier 40. A secondary-side switching current I_(S) is generated accordingly. A peak value I_(S1) of the secondary-side switching current I_(S) is expressed by the following:

$\begin{matrix} {I_{S\; 1} = {\frac{\left( {V_{O} + V_{F}} \right)}{L_{S}} \times T_{DS}}} & (2) \end{matrix}$ where V_(O) is the output voltage of the switching regulator, V_(F) is a forward voltage drop of the rectifier 40, and L_(S) is the inductance of the secondary winding N_(S) of the transformer 10, and T_(DS) is a discharge time of the secondary-side switching current I_(S).

Meanwhile, a reflected signal V_(AUX) is generated at the auxiliary winding N_(A) of the transformer 10. The reflected signal V_(AUX) is given by:

$\begin{matrix} {V_{AUX} = {\frac{T_{NA}}{T_{NS}} \times \left( {V_{O} + V_{F}} \right)}} & (3) \end{matrix}$ where T_(NA) and T_(NS) are a plurality of winding turns of the auxiliary winding N_(A) and the secondary winding N_(S) of the transformer 10, respectively.

The reflected signal V_(AUX) is started to decrease as the secondary-side switching current I_(S) is fallen to zero. This is also indicative that the energy of the transformer 10 is fully released at the moment. Therefore, as shown in FIG. 2, the discharge time T_(DS) in equation (2) is measured from the falling edge of the switching signal V_(PWM) to the point that the reflected signal V_(AUX) is started to fall. The secondary-side switching current I_(S) is determined by the primary-side switching current I_(P) and the winding turns of the transformer 10. The secondary-side switching current I_(S) is expressed by:

$\begin{matrix} {I_{s} = {\frac{T_{NP}}{T_{NS}} \times I_{P}}} & (4) \end{matrix}$ where T_(NP) is a plurality of winding turns of the primary winding N_(P) of the transformer 10.

Referring to FIG. 1, the control circuit 70 includes a supply terminal VCC and a ground terminal GND for receiving power. A divider connected between the auxiliary winding N_(A) of the transformer 10 and a ground reference level is formed by a resistor 50 and a resistor 51. A detection terminal DET of the control circuit 70 is connected to a joint of the resistor 50 and the resistor 51. A voltage V_(DET) generated at the detection terminal DET is given by:

$\begin{matrix} {V_{DET} = {\frac{R_{51}}{R_{50} + R_{51}} \times V_{AUX}}} & (5) \end{matrix}$ where R₅₀ and R₅₁ are the resistance of the resistors 50 and 51.

A capacitor 65 is further charged by the reflected signal V_(AUX) via a rectifier 60 to power the control circuit 70. A current-sense resistor 30 is served as a current sense device. The current-sense resistor 30 is connected from a source of the transistor 20 to the ground reference level for converting the primary-side switching current I_(P) into a current-sense signal V_(CS). A sense terminal CS of the control circuit 70 is connected to the current-sense resistor 30 for detecting the current-sense signal V_(CS).

The switching signal V_(PWM) is generated by an output terminal OUT of the control circuit 70 to switch the transformer 10. A voltage-compensation terminal COMV is connected to a first compensation network for frequency compensation for the first error amplifier. The first compensation network can be a capacitor connected to the ground reference level, such as a capacitor 31. A current-compensation terminal COMI has a second compensation network for frequency compensation for the second error amplifier. The second compensation network can also be a capacitor connected to the ground reference level, such as a capacitor 32. A programmable terminal COMR has a resistor 33 connected to ground to adjust a reference signal V_(REF) for a voltage feedback loop of the control circuit 70 in accordance with the output current I_(O). The adjustment of the reference signal V_(REF) is to compensate the voltage drop of the output cable 46 for achieving better load regulation.

FIG. 3 illustrates the control circuit 70 according to an embodiment of the present invention. A first signal V_(V) and a timing signal S_(DS) is produced by a first circuit 100 by multi-sampling the voltage V_(DET). The discharge time T_(DS) of the secondary-side switching current I_(S) is represented by the timing signal S_(DS). A current signal V_(W) is generated by a second circuit 300 by measuring the current-sense signal V_(CS). An oscillation signal PLS is generated by an oscillator 200 for determining a switching frequency of the switching signal V_(PWM). A second signal V_(I) is produced by a third circuit 400 by integrating the current signal V_(W) with the timing signal S_(DS). A first error amplifier is developed by an operational amplifier 71 and a reference signal V_(REF) for amplifying the first signal V_(V) and providing a first feedback loop circuit for output voltage control. A second error amplifier is developed by an operational amplifier 72 and a reference signal V_(REF2) for amplifying the second signal V_(I) and providing a second feedback loop circuit for output current control. An adjust circuit 700 is coupled to the programmable terminal COMR to adjust the reference signal V_(REF) in accordance with a reference signal V_(REF1) and the second signal V_(I). A PWM circuit 500 and a plurality of comparators 73 and 75 form a switching control circuit to generate the switching signal V_(PWM) and to control the pulse width of the switching signal V_(PWM) in response to the outputs of the first error amplifier and the second error amplifier. Both of the operational amplifiers 71 and 72 have trans-conductance output. The output of the operational amplifier 71 is connected to the voltage-compensation terminal COMV and a positive input of the comparator 73. The output of the operational amplifier 72 is connected to the current-compensation terminal COMI and a positive input of the comparator 75. A negative input of the comparator 73 is connected to an output of an adder 600. A negative input of the comparator 75 is supplied with a ramp signal RMP that is produced from the oscillator 200.

A slope signal V_(SLP) is generated by the adder 600 by adding the current-sense signal V_(CS) with the ramp signal RMP. A positive input of a comparator 74 is supplied with a reference signal V_(REF3). A negative input of the comparator 74 is connected to the sense terminal CS for achieving a cycle-by-cycle current limit. Three inputs of a NAND gate 79 are respectively connected to the outputs of the comparators 73, 74 and 75. A reset signal RST is generated by an output of the NAND gate 79. The reset signal RST is supplied to the PWM circuit 500 for controlling the duty cycle of the switching signal V_(PWM).

The current control loop is formed from the detection of the primary-side switching current I_(P) to the pulse width modulation of the switching signal V_(PWM) to control the magnitude of the primary-side switching current I_(P) in response to the reference signal V_(REF2). The secondary-side switching current I_(S) is a ratio of the primary-side switching current I_(P) as shown in equation (4). According to the signal waveforms in FIG. 2, the output current I_(O) of the switching regulator is the average of the secondary-side switching current I_(S). It is expressed by the following:

$\begin{matrix} {I_{O} = {I_{S} \times \frac{T_{DS}}{2T}}} & (6) \end{matrix}$ Therefore, the output current (I_(O)) of the switching regulator is regulated.

The current-sense signal V_(CS) is detected by the second circuit 300; and the current signal V_(W) is generated. The second signal V_(I) is further produced by the third circuit 400 by integrating the current signal V_(W) with the discharge time T_(DS). The second signal V_(I) is thus designed as the following:

$\begin{matrix} {V_{I} = {\frac{V_{W}}{2} \times \frac{T_{DS}}{T_{I}}}} & (7) \end{matrix}$ where the current signal V_(W) is expressed by the following:

$\begin{matrix} {V_{W} = {\frac{T_{NS}}{T_{NP}} \times R_{S} \times I_{S}}} & (8) \end{matrix}$ where T_(I) is a time constant of the third circuit 400. It is seen from equations (6)-(8), the second signal V_(I) is rewritten as the following:

$\begin{matrix} {V_{I} = {\frac{T}{T_{I}} \times \frac{T_{NS}}{T_{NP}} \times R_{S} \times I_{O}}} & (9) \end{matrix}$

It is found that the second signal V_(I) is proportional to the output current I_(O) of the switching regulator. The second signal V_(I) is increased as the output current I_(O) increases. However, the maximum value of the second signal V_(I) is limited to the value of the reference signal V_(REF2) through the regulation of the current control loop. Under the feedback control of the current control loop, a maximum output current I_(O(max)) is given by the following:

$\begin{matrix} {I_{O{(\max)}} = {\frac{T_{NP}}{T_{NS}} \times \frac{G_{A} \times G_{SW} \times V_{{REF}\; 2}}{1 + \left( {G_{A} \times G_{SW} \times \frac{R_{S}}{K}} \right)}}} & (10) \end{matrix}$ where K is a constant equal to T_(I)/T, G_(A) is the gain of the second error amplifier, and G_(SW) is the gain of the switching circuit.

As the loop gain of the current control loop is high (G_(A)×G_(SW)>>1), the maximum output current I_(O(max)) could be briefly defined as the following:

$\begin{matrix} {I_{O{(\max)}} = {K \times \frac{T_{NP}}{T_{NS}} \times \frac{V_{{REF}\; 2}}{R_{S}}}} & (11) \end{matrix}$

The maximum output current I_(O(max)) of the switching regulator is thus regulated as a constant current in response to the reference signal V_(REF2).

Besides, a voltage control loop is developed from the sampling of the reflected signal V_(AUX) to the pulse width modulation of the switching signal V_(PWM), which the magnitude of the reflected signal V_(AUX) is controlled in response to the reference signal V_(REF). The reflected signal V_(AUX) is a ratio of the output voltage V_(O) as shown in equation (3). The reflected signal V_(AUX) is further attenuated to the voltage V_(DET) as shown in equation (5). The first signal V_(V) is generated by the first circuit 100 by multi-sampling the voltage V_(DET). The value of the first signal V_(V) is controlled in response to the value of the reference signal V_(REF) through the regulation of the voltage control loop. The loop gain for the voltage control loop is provided by the first error amplifier and the switching circuit. Therefore, the output voltage V_(O) is briefly defined as the following:

$\begin{matrix} {V_{O} = {\left( {\frac{R_{50} + R_{51}}{R_{50}} \times \frac{T_{NS}}{T_{NA}} \times V_{REF}} \right) - V_{F}}} & (12) \end{matrix}$

The reflected signal V_(AUX) is multi-sampled by the first circuit 100. The voltage is sampled and measured instantly before the secondary-side switching current I_(S) is fallen to zero. Therefore, the variation of the secondary-side switching current I_(S) does not affect the value of the forward voltage drop V_(F) of the rectifier 40. However, the voltage drop of the output cable is varied when the output current I_(O) is changed. The adjust circuit 700 is developed to compensate the voltage drop of the output cable. The resistor 33 is utilized to program a slope to determine the change of the reference signal V_(REF) in response to the change of the second signal V_(I). Therefore, the voltage drop is compensated in proportional to the output current I_(O). Using different value of the resistor 33, the compensation can be programmed for a variety of the output cables 46.

FIG. 4 illustrates the first circuit 100 according to an embodiment of the present invention. A sample-pulse signal is produced by a sample-pulse generator 190 for multi-sampling. A threshold voltage 156 is added up with the reflected signal V_(AUX) to produce a level-shift reflected signal. A first signal generator includes a counter 171, and the AND gates 165 . . . 166 for producing the sample signals V_(SP1) . . . V_(SPN). A second signal generator includes a D flip-flop 170, an NAND gate 163, an AND gate 164, and a comparator 155 for producing the timing signal S_(DS). A time-delay circuit includes an inverter 162, a current source 180, a transistor 181, and a capacitor 182 for generating a delay time T_(d) as the switching signal V_(PWM) is disabled. An input of an inverter 161 is supplied with the switching signal V_(PWM). An output of the inverter 161 is connected to an input of the inverter 162, a first input of the AND gate 164, and a clock-input of the D flip-flop 170. The transistor 181 is turned on/off by an output of the inverter 162. The capacitor 182 is connected in parallel with the transistor 181. The current source 180 is applied to charge the capacitor 182. Therefore, the delay time T_(d) of the time-delay circuit is determined by the current of the current source 180 and the capacitance of the capacitor 182. An output of the time-delay circuit is obtained across the capacitor 182. A D-input of the D flip-flop 170 is pulled high by a supply voltage V_(CC). An output of the D flip-flop 170 is connected to a second input of the AND gate 164. The timing signal S_(DS) is outputted by the AND gate 164. The timing signal S_(DS) is thus enabled as the switching signal V_(PWM) is disabled. The output of the NAND gate 163 is connected to a reset-input of the D flip-flop 170. Two inputs of the NAND gate 163 are respectively connected to the output of the time-delay circuit and the output of the comparator 155. A negative input of the comparator 155 is supplied with the level-shift reflected signal. A positive input of the comparator 155 is supplied with a hold voltage V_(HD). Therefore, after the delay time T_(d), the timing signal S_(DS) is disabled once the level-shift reflected signal is lower than the hold voltage V_(HD). Besides, the timing signal S_(DS) is also disabled as long as the switching signal V_(PWM) is enabled.

The sample-pulse signal is supplied to the counter 171 and the third inputs of the AND gates 165 . . . 166. The outputs of the counter 171 are respectively connected to the second inputs of the AND gates 165 . . . 166. The first inputs of the AND gates 165 . . . 166 are supplied with the timing signal S_(DS). The fourth inputs of the AND gates 165 . . . 166 are connected to the output of the time-delay circuit. Therefore, the sample signals V_(SP1) . . . V_(SPN) are generated in response to the sample-pulse signal. Besides, the sample signals V_(SP1) . . . V_(SPN) are alternately produced during an enabled period of the timing signal S_(DS). However, the delay time T_(d) is inserted at the beginning of the timing signal S_(DS) to inhibit the sample signals V_(SP1) . . . V_(SPN). The sample signals V_(SP1) . . . V_(SPN) are thus disabled during the period of the delay time T_(d).

The sample signals V_(SP1) . . . V_(SPN) are used for sampling the reflected signal V_(AUX) in sequence via the detection terminal DET and the divider. The switches 121 . . . 122 are controlled by the sample signals V_(SP1) . . . V_(SPN) for obtaining the hold voltages across the capacitors 110 . . . 111, respectively. The switches 123 . . . 124 are connected in parallel with the capacitors 110 . . . 111 to discharge the capacitors 110 . . . 111. A buffer circuit includes the operational amplifiers 150 . . . 151, the diodes 130 . . . 131, and a current source 135 for generating the hold voltage V_(HD). The positive inputs of the operational amplifiers 150 . . . 151 are connected to the capacitors 110 . . . 111, respectively. The negative inputs of the operational amplifiers 150 . . . 151 are connected to an output of the buffer circuit. The diodes 130 . . . 131 are connected from the output of the operational amplifiers 150 . . . 151 to the output of the buffer circuit. The hold voltage V_(HD) is thus obtained from the higher voltage of hold voltages. The current source 135 is used for the termination. The hold voltage V_(HD) is periodically conducted by a switch 125 to a capacitor 115 for producing the first signal V_(V). The switch 125 is turned on/off via the oscillation signal PLS. After the delay time T_(d), the sample signals V_(SP1) . . . V_(SPN) are started to produce hold voltages. The spike interference of the reflected signal V_(AUX) are eliminated. The spike interference of the reflected signal V_(AUX) would occur when the switching signal V_(PWM) is disabled and the transistor 20 is turned off.

As the secondary-side switching current I_(S) falls to zero, the reflected signal V_(AUX) is started to decrease. The aforementioned shall be detected by the comparator 155 to disable the timing signal S_(DS). Therefore, the pulse width of the timing signal S_(DS) is correlated to the discharge time T_(DS) of the secondary-side switching current I_(S). Meanwhile, the sample signals V_(SP1) . . . V_(SPN) are disabled, and the multi-sampling is stopped as the timing signal S_(DS) is disabled. At the moment, the hold voltage V_(HD) generated at the output of the buffer circuit is thus correlated to the reflected signal V_(AUX) that is sampled once the secondary-side switching current I_(S) is fallen to zero. The hold voltage V_(HD) is obtained from the higher voltage of hold voltages, which shall ignore the voltage that is sampled when the reflected signal had started to decrease.

FIG. 5 illustrates the oscillator 200 according to an embodiment of the present invention. A first V-to-I converter is formed by an operational amplifier 201, a resistor 210, and a transistor 250. A reference current I₂₅₀ is generated by the first V-to-I converter in response to a reference signal V_(R). Current mirrors are formed from a plurality of transistors, for example, transistors 251, 252, 253, 254 and 255 for generating an oscillator charge current I₂₅₃ and an oscillator discharge current I₂₅₅ in response to the reference current I₂₅₀. The oscillator charge current I₂₅₃ is generated by a drain of the transistor 253. The oscillator discharge current I₂₅₅ is generated by a drain of the transistor 255. A switch 230 is connected between the drain of the transistor 253 and a capacitor 215. A switch 231 is connected between the drain of the transistor 255 and the capacitor 215. The ramp signal RMP is obtained across the capacitor 215. A comparator 205 has a positive input connected to the capacitor 215. The oscillation signal PLS is outputted by the comparator 205. The switching frequency of the switching signal V_(PWM) is determined by the oscillation signal PLS. A first terminal of a switch 232 is supplied with a high threshold voltage V_(H). A first terminal of a switch 233 is supplied a low threshold voltage V_(L). A second terminal of the switch 232 and a second terminal of the switch 233 are both connected to a negative input of the comparator 205. An input of an inverter 260 is connected to an output of the comparator 205 for producing an inverse oscillation signal /PLS. The switch 231 and the switch 233 are turned on/off by the oscillation signal PLS. The switch 230 and the switch 232 are turned on/off by the inverse oscillation signal /PLS. The resistance R₂₁₀ of the resistor 210 and the capacitance C₂₁₅ of the capacitor 215 are to determine a switching period T of the switching frequency:

$\begin{matrix} {T = {\frac{C_{215} \times V_{OSC}}{V_{R}/R_{210}} = {R_{210} \times C_{215} \times \frac{V_{OSC}}{V_{R}}}}} & (13) \end{matrix}$

where V_(OSC)=V_(H)−V_(L).

FIG. 6 illustrates the second circuit 300 according to an embodiment of the present invention. A fourth circuit includes a comparator 310, a current source 320, a plurality of switches 330, 340, and a capacitor 361. The peak value of the current-sense signal V_(CS) is sampled for generating a fourth signal. A positive input of the comparator 310 is supplied with the current-sense signal V_(CS). A negative input of the comparator 310 is connected to the capacitor 361. The switch 330 is connected between the current source 320 and the capacitor 361. The switch 330 is turned on/off by an output of the comparator 310. The switch 340 is connected in parallel with the capacitor 361 for discharging the capacitor 361. The fourth signal is periodically conducted by a switch 350 to a capacitor 362 for producing the current signal V_(W). The switch 350 is turned on/off by the oscillation signal PLS.

FIG. 7 illustrates the third circuit 400 according to an embodiment of the present invention. A second V-to-I converter includes an operational amplifier 410, a resistor 450, and the transistors 420, 421, 422. A positive input of the operational amplifier 410 is supplied with the current signal V_(W). A negative input of the operational amplifier 410 is connected to the resistor 450. A gate of the transistor 420 is driven by an output of the operational amplifier 410. A source of the transistor 420 is coupled to the resistor 450. A current I₄₂₀ is generated by the second V-to-I converter via a drain of the transistor 420 in response to the current signal V_(W). A current mirror having a 2:1 ratio is formed by the transistors 421 and 422. The current mirror is driven by the current I₄₂₀ to produce a programmable charge current I_(PRG) via a drain of the transistor 422. The programmable charge current I_(PRG) is expressed by the following:

$\begin{matrix} {I_{PRG} = {\frac{1}{R_{450}} \times \frac{V_{W}}{2}}} & (14) \end{matrix}$ where R₄₅₀ is the resistance of the resistor 450.

A capacitor 471 is used to produce an integrated signal. A switch 460 is connected between the drain of the transistor 422 and the capacitor 471. The switch 460 is turned on/off by the timing signal S_(DS). A switch 462 is connected in parallel with the capacitor 471 for discharging the capacitor 471. The integrated signal is periodically conducted by a switch 461 to a capacitor 472 for producing the second signal V_(I). The switch 461 is turned on/off by the oscillation signal PLS. The second signal V_(I) is therefore obtained across the capacitor 472 as shown:

$\begin{matrix} {V_{I} = {\frac{1}{R_{450} \times C_{471}} \times \frac{V_{W}}{2} \times T_{DS}}} & (15) \end{matrix}$

According to an embodiment of the present invention illustrated in FIG. 4˜7, the second signal V_(I) is correlated to the secondary-side switching current I_(S) and the output current I_(O) of the switching regulator. Thus, the equation (9) is rewritten as the following:

$\begin{matrix} {V_{I} = {m \times \frac{T_{NS}}{T_{NP}} \times R_{S} \times I_{O}}} & (16) \end{matrix}$ where m is a constant, which is determined by the following:

$\begin{matrix} {m = {\frac{R_{210} \times C_{216}}{R_{450} \times C_{471}} \times \frac{V_{OSC}}{V_{R}}}} & (17) \end{matrix}$

The resistance R₄₅₀ of the resistor 450 is correlated to the resistance R₂₁₀ of the resistor 210. The capacitance C₄₇₁ of the capacitor 471 is correlated to the capacitance C₂₁₅ of the capacitor 215. Therefore, the second signal V_(I) is proportional to the output current I_(O) of the switching regulator.

FIG. 8 illustrates a circuit schematic of the PWM circuit 500 according to an embodiment of the present invention. The PWM circuit 500 includes a NAND gate 511, a D flip-flop 515, an AND gate 519, a blanking circuit 520, and the inverters 512, 518. A D-input of the D flip-flop 515 is pulled high by the supply voltage V_(CC). An input of the inverter 512 is driven by the oscillation signal PLS. An output of the inverter 512 is connected to a clock-input of the D flip-flop 515 for enabling the switching signal V_(PWM). An output of the D flip-flop 515 is connected to a first input of the AND gate 519. A second input of the AND gate 519 is coupled to the output of the inverter 512. The switching signal V_(PWM) is outputted by the AND gate 519 to switch the transformer 10. A reset-input of the D flip-flop 515 is connected to an output of the NAND gate 511. A first input of the NAND gate 511 is supplied with the reset signal RST for the cycle-by-cycle disabling of the switching signal V_(PWM). The second input of the NAND gate 511 is connected to an output of the blanking circuit 520 for ensuring a minimum on-time of the switching signal V_(PWM) when the switching signal V_(PWM) is enabled. A minimum value of the discharge time T_(DS) is ensured by the minimum on-time of the switching signal V_(PWM), which a proper multi-sampling of the reflected signal V_(AUX) in the first circuit 100 is ensured. The discharge time T_(DS) is correlated to the on-time of the switching signal V_(PWM). Referring to equations (1), (2), and (4), and the secondary inductance L_(S), which is shown in equation (18), the discharge time T_(DS) is expressed as equation (19) as shown in the following:

$\begin{matrix} {L_{S} = {\left( {T_{NS}/T_{NP}} \right)^{2} \times L_{P}}} & (18) \\ {T_{DS} = {\left( \frac{V_{IN}}{V_{O} + V_{F}} \right) \times \frac{T_{NS}}{T_{NP}} \times T_{ON}}} & (19) \end{matrix}$ where T_(ON) is the on-time of the switching signal V_(PWM).

An input of the blanking circuit 520 is supplied with the switching signal V_(PWM). When the switching signal V_(PWM) is enabled, the blanking circuit 520 shall generate a blanking signal V_(BLK) to inhibit the reset of the D flip-flop 515. The blanking circuit 520 further includes an NAND gate 523, a current source 525, a capacitor 527, a transistor 526, and a plurality of inverters 521, 522. The switching signal V_(PWM) is supplied to an input of the inverter 521 and the first input of the NAND gate 523. The current source 525 is applied to charge the capacitor 527. The capacitor 527 is connected in parallel with the transistor 526. The transistor 526 is turned on/off by an output of the inverter 521. An input of the inverter 522 is coupled to the capacitor 527. An output of the inverter 522 is connected to a second input of the NAND gate 523. The blanking signal V_(BLK) is outputted by an output of the NAND gate 523. The pulse width of the blanking signal V_(BLK) is determined by the current of the current source 525 and the capacitance of the capacitor 527. An input of an inverter 518 is connected to the output of the NAND gate 523. A clear signal (CLR) is generated by an output of the inverter 518 to turn on/off the switches 123, 124, 340 and 462.

FIG. 9 illustrates a circuit schematic of the adder 600 according to an embodiment of the present invention. A third V-to-I converter is formed by an operational amplifier 610, a plurality of transistors 620, 621, 622, and a resistor 650 for generating a current I₆₂₂ in response to the ramp signal RMP. A positive input of an operational amplifier 611 is supplied with the current-sense signal V_(CS). A negative input and an output of the operational amplifier 611 are connected together to build the operational amplifier 611 as a buffer. A drain of the transistor 622 is connected to the output of the operational amplifier 611 via a resistor 651. The slope signal V_(SLP) is generated at the drain of the transistor 622. The slope signal V_(SLP) is therefore correlated to the ramp signal RMP and the current-sense signal V_(CS).

FIG. 10 illustrates a circuit schematic of the adjust circuit 700 according to an embodiment of the present invention. A V-to-I converter is formed by an operational amplifier 710, a plurality of transistors 711, 714, 715, and a resistor 712 for generating a current I₇₁₅ in response to the second signal V_(I). A positive input of an operational amplifier 710 is supplied with the second signal V_(I). The current I₇₁₅ is outputted to the programmable terminal COMR. The current I₇₁₅ is associated with the resistor 33 which generates a voltage V_(COMR) connected to an operational amplifier 720. Another V-to-I converter is formed by the operational amplifier 720, a plurality of transistors 721, 724, 725, and a resistor 722 for generating a current I₇₂₅ at the drain of the transistor 725 in response to the voltage V_(COMR). A negative input and an output of the operational amplifier 750 are connected together to build the operational amplifier 750 as a buffer. The positive input of the operational amplifier 750 is connected to the reference signal V_(REF1). A drain of the transistor 725 is connected to the output of the operational amplifier 750 via a resistor 760. The reference signal V_(REF) is generated at the drain of the transistor 725. Based on the reference signal V_(REF1), the reference signal V_(REF) is adjusted by the second signal V_(I) and is programmed by the resistor 33.

It will be apparent to those skilled in the art that various modifications and variations can be made to the structure of the present invention without departing from the scope or spirit of the invention. In view of the foregoing, it is intended that the present invention covers modifications and variations of this invention provided they fall within the scope of the following claims and their equivalents. 

1. A switching regulator, comprising: a transformer, having a primary side and a secondary side; a switching device, for switching the transformer; a control circuit, coupled to the transformer for generating a switching signal to switch the switching device and to regulate the output of the switching regulator; wherein the control circuit comprises: a first circuit, coupled to the transformer for generating a first signal in response to a reflected signal of the transformer; a second circuit, generating a second signal in response to a primary-side switching current of the transformer; a feedback circuit, generating a feedback signal in response to the first signal, and the feedback signal is varied in response to the second signal; and a switching control circuit, generating the switching signal in response to the feedback signal.
 2. The switching regulator as claimed in claim 1, wherein the first circuit comprises: a plurality of capacitors; a signal generator producing sample signals to sample and to hold the reflected signal to the capacitors, wherein hold voltages are respectively generated across the capacitors, and samples signals are generated in response to the enable of a timing signal; a buffer circuit, generating the first signal in accordance with the hold voltages; a second signal generator, producing the timing signal in response to hold voltage, wherein the timing signal is enabled as the switching signal is disabled, and the timing signal is disabled when the reflected signal is significant lower than the hold signal.
 3. The switching regulator as claimed in claim 1, wherein the first circuit multi-samples the reflected signal for generating the first signal, and the first signal is acquired once the discharge current of the transformer is fallen to zero.
 4. The switching regulator as claimed in claim 1, wherein the second circuit comprises: a current generator, generating a charge current in response to an output current of the switching regulator; and a capacitor, coupled to the charge current for generating the second signal. 